Receiver system with interdependent adaptive analog and digital signal equalization

ABSTRACT

An analog equalizer ( 613  and  614 ) adaptively equalizes an input analog signal affected with intersymbol interference (“ISI”), or an intermediate analog signal generated therefrom, to produce a filtered partially equalized analog signal with reduced ISI. An analog-to-digital converter ( 210 ) converts the filtered analog signal, or an intermediate analog signal generated therefrom, into an initial digital signal. A digital equalizer ( 212 ) adaptively equalizes the initial digital signal, or an intermediate digital signal generated therefrom, to produce an equalized digital signal as a stream of equalized digital values with further reduced ISI. An output decoder ( 605 ) decodes the equalized digital values, or intermediate digital values generated therefrom, into a stream of symbols. Equalization control circuitry ( 213, 214,  and  217 ) adjusts equalization filter characteristics of the equalizers such that adjustments of the filter characteristics of one of the equalizers depend adaptively on adaptive adjustments of the filter characteristics of the other equalizer.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is a continuation of U.S. patent application Ser. No. 11/490,437,filed 19 Jul. 2006, now U.S. Pat. No. 7,646,807 B1, which is a divisionof U.S. patent application Ser. No. 09/561,086, filed Apr. 28, 2000, nowU.S. Pat. No. 7,254,198 B1.

FIELD OF THE INVENTION

This invention relates to digital communication systems and, moreparticularly, to an optimal architecture for receiver processing.

BACKGROUND

The dramatic increase in desktop computing power driven byintranet-based operations and the increased demand for time-sensitivedelivery between users has spurred development of high speed Ethernetlocal area networks (LANs). 100BASE-TX Ethernet (see IEEE Std.802.3u-1995 CSMA/CD Access Method, Type 100 Base-T) using existingcategory 5 (CAT-5) copper wire, and the newly developing 1000BASE-TEthernet (see IEEE Draft P802.3ab/D4.0 Physical Layer Specification for1000 Mb/s Operation on Four Pairs of Category 5 or Better Twisted PairCable (1000 Base-T)) for gigabit-per-second transfer of data overcategory 5 data grade copper wiring, require new techniques in highspeed symbol processing. On category 5 cabling, gigabit-per-secondtransfer can be accomplished utilizing four twisted pairs and a 125megasymbol-per-second transfer rate on each pair where each symbolrepresents two bits.

Physically, data is transferred using a set of voltage pulses where eachvoltage represents one or more bits of data. Each voltage in the set isreferred to as a symbol and the whole set of voltages is referred to asa symbol alphabet.

One system of transferring data at high rates is Non-Return-to-Zero(NRZ) signaling. In NRZ, the symbol alphabet {A} is {−1, +1}. A logical“1” is transmitted as a positive voltage while a logical “0” istransmitted as a negative voltage. At 125 megasymbols per second, thepulse width of each symbol (the positive or negative voltage) is 8 ns.

An alternative modulation method for high speed symbol transfer isMultilevel Threshold-3 (MLT3) and involves a three-level system. (SeeAmerican National Standard Information System, Fibre Distributed DataInterface (FDDI)-Part: Token Ring Twisted Pair Physical Layer MediumDependent (TP-PMD), ANSI X3.263:199X.) The symbol alphabet {A} for MLT3is {−1, 0, +1}. In MLT3 transmission, a logical “1” is transmitted byeither a −1 or a +1 while a logical “0” is transmitted as a 0. Atransmission of two consecutive logical “1”s does not require the systemto pass through zero in the transition. A transmission of the logicalsequence (“1”, “0”, “1”) results in transmission of the symbols (+1, 0,−1) or (−1, 0, +1), depending on the symbols transmitted prior to thissequence. If the symbol transmitted immediately prior to the sequencewas a +1, the symbols (+1, 0, −1) are transmitted. If the symboltransmitted before this sequence was a −1, the symbols (−1, 0, +1) aretransmitted. If the symbol transmitted immediately before this sequencewas a 0, the first symbol of the sequence transmitted will be a +1 ifthe previous logical “1” was transmitted as a −1 and will be a −1 if theprevious logical “1” was transmitted as a +1.

The detection system in the MLT3 standard, however, needs to distinguishbetween three levels, instead of two levels as in a more typicaltwo-level system. The signal-to-noise ratio required to achieve aparticular bit error rate is higher for MLT3 signaling than fortwo-level systems. The advantage of the MLT3 system, however, is thatthe energy spectrum of the emitted radiation from the MLT3 system isconcentrated at lower frequencies and therefore more easily meets FCCradiation emission standards for transmission over twisted pair cables.Other communication systems may use a symbol alphabet having more thantwo voltage levels in the physical layer in order to transmit multiplebits of data using each individual symbol. In Gigabit Ethernet overtwisted pair CAT-5 cabling, for example, five-level Pulse-AmplitudeModulation (PAM-5) data can be transmitted at a baud rate of 125megabaud. (See IEEE Draft P802.3ab/D4.0 Physical Layer Specification for1000 Mb/s Operation on Four Pairs of Category 5 or Better Twisted PairCable (1000 Base-T).)

Therefore, there is a necessity for a receiver capable of receivingsignals having large intersymbol interference from long transmissioncables. There is also a necessity for reducing the difficultiesassociated with digital equalization of signals with large intersymbolinterference without losing the equalization versatility required tooptimize the receiver.

SUMMARY OF THE INVENTION

In accordance with the invention, a receiver system for providing signalequalization is partitioned into an analog pre-filter and a digitalreceiver. At least some of the intersymbol interference is removed fromthe signal by the analog pre-filter before the signal is processedthrough a digital equalizer in the digital receiver. Signals having alarge amount of intersymbol interference, such as those transmittedthrough long cables, are preprocessed through the pre-filter, therebyreducing the difficulties of digital equalization without losing theversatility of the digital equalizer.

Embodiments of the invention can include any equalization scheme,including linear equalization, decision feedback equalization, trellisdecoding and sequence decoding, separately or in combination.Embodiments of the invention may also include cable quality and cablelength indication and baseline wander correction. Further, embodimentsof receivers according to the present invention can also include echocancellation and near end crosstalk (NEXT) cancellation.

These and other embodiments of the invention are further explained belowalong with the following figures.

DESCRIPTION OF THE FIGURES

FIG. 1A shows a transmission system with an entirely digital equalizerin the receiver.

FIG. 1B shows a transmission system with an entirely analog equalizer inthe receiver.

FIG. 2A shows a transmission system that contains a receiver systemaccording to the present invention.

FIG. 2B shows an analog model of a transmission channel.

FIG. 3A shows an exemplary transfer function representing a transmissionchannel.

FIG. 3B shows the exponential component of the signal distortion.

FIG. 3C shows an exemplary transfer function of a pre-filter accordingto the present invention.

FIG. 3D shows the combined influence of the functions shown in FIGS. 3A,3B and 3C on an input signal.

FIG. 4 shows a discrete time model of signal transmission through atransmission channel in combination with a pre-filter according to thepresent invention.

FIG. 5A shows another embodiment of a receiver according to the presentinvention.

FIGS. 5B and 5C show embodiments of pre-filters for a receiver accordingto the present invention.

FIGS. 6A, 6B, 6C, 6D and 6E show embodiments of a multi-wire receiversystem according to the present invention.

In the figures, elements having similar or identical functions may haveidentical identifiers.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1A shows a block diagram of a typical transmission system 100 for asingle-wire transmission channel. Transmission system 100 includestransmitter 107, transmission channel 102 and receiver 103. Transmitter107 transmits a symbol stream {a_(k)} and can perform some signalshaping on the waveform formed by symbol stream {a_(k)}. Notationally, aparticular symbol in clock cycle k is denoted without brackets as a_(k)whereas the symbol sequence or stream is denoted with curly brackets as{a_(k)}.

Transmission channel 102 which can be any transmission medium distortsthe transmitted waveform, creates intersymbol interference, and addsnoise to the transmitted signal. Receiver 103 receives the transmittedsignals from transmission channel 102. Receiver 103 includes ananalog-to-digital converter (ADC) 104 and an equalizer 106 connected inseries. In receiver 103 of FIG. 1A, the equalization of an input signalto receiver 103 is accomplished digitally. Digital equalization becomesproblematic as the cable length increases due to the large intersymbolinterference associated with longer cables.

In general, a signal received by receiver 103 includes contributionsfrom several transmitted symbols as well as noise and channeldistortions. Each transmitted symbol is diffused in the transmissionprocess so that it is commingled with symbols being transmitted at latertransmission times. This effect is known as “intersymbol interference”(ISI). (See E. A. LEE AND D. G. MESSERCHMITT, DIGITAL COMMUNICATIONS(1988).)

Intersymbol interference is a result of the dispersive nature of thecommunication channel. The IEEE LAN standards require that LANcommunication systems be capable of transmitting and receiving datathrough at least a 100 meter cable. In a 100 meter cable, the signalstrength at the Nyquist frequency of 62.5 Mhz is reduced nearly 20 db atthe receiving end of the cable. Given this dispersion, a single symbolmay affect symbols throughout the transmission cable of transmissionchannel 102.

An input signal x_(k) to receiver 103 at sample time k, neglectingchannel distortion and noise, can be digitally represented asx _(k) =C ₀ a _(k) +C ₁ a _(k−1) + . . . +C _(j) a _(k−j) . . . ,  (1)where a_(k−j) represents the (k−j)th symbol in the symbol sequence andcoefficient C_(j) represents the contribution of the (k−j)th symbol tosignal x_(k). Equalizer 106 receives digitized sample x_(k) and deducescurrently received symbol â_(k) by removing, usually adaptively, thecontribution of previous symbols a_(k−j) from detected sample x_(k)(i.e., by removing the intersymbol interference). The deduced symbolâ_(k) represents the best estimation by receiver 103 as to what thetransmitted symbol a_(k) was.

However, with long cable lengths, the contribution of earlier receivedsymbols becomes significant. For example, with cable lengths above about100 meters, coefficient C₁ for immediately previous symbol a_(k−1) canbe as high as 0.95 (i.e., 95% of symbol a_(k−1) may be represented inthe input signal). Contributions from other previous symbols can also behigh. Given that equalizer 106 cannot adjust for the contribution ofsymbols not yet received (e.g., the kth detected sample cannot includecontributions from the (k+1)th transmitted symbol), equalizer 106 has adifficult time distinguishing the kth and the (k−1)th symbol under thesecircumstances. An adaptive receiver can have particular difficulty uponstartup in distinguishing the contribution of the kth symbol from thecontribution of the (k−1)th symbol and in determining the equalizerparameters corresponding to the mixing parameters {C_(j)}.

Therefore, for large cable lengths a digital equalizer is faced withdeducing the current symbol from a sample containing significantcontributions from numerous previously received symbols. The difficultyis not only deducing the symbols but in adaptively choosing theoperating parameters of the equalizer in order to optimize theperformance of receiver 103.

An alternative approach to digital equalization is analog equalization.FIG. 1B shows a block diagram of a transmission system 110 having ananalog receiver 111 coupled to transmission channel 102. Receiver 111includes an analog equalizer 112 having an analog filter tailored toremove intersymbol interference from the received signal. Althoughhaving the advantage of processing loop speed, equalizer 112 cannot beadaptively optimized for the performance of receiver 111.

FIG. 2A shows a transmission system 200 according to the presentinvention. Transmission system 200 includes a transmitter 221, atransmission channel 201 and a receiver system 206. Transmitter 221outputs a symbol stream {a_(k)} to transmission channel 201. Transmitter221 can output symbol stream {a_(k)} directly or, in some embodiments ofthe invention, perform some pre-processing of the waveform formed by thesequential transmission of the symbol stream {a_(k)}.

Transmission channel 201 represents the transmission of a signal betweentransmitter 221 and receiver 206 and can include any transmissionmedium, including twisted copper, coaxial cable or optical fiber. Thesymbol stream {a_(k)} can be composed of any symbol alphabet, includingNRZ, MLT3, PAM-5 (where the symbol alphabet is given by {−2, −1, 0, 1,2}) or any other symbol alphabet and modulation that are used intransceivers such as transmission system 200.

Transmission system 200 may be a portion of a larger transceiver system.In general, transceivers of this type may have any number oftransmission channels similar to transmission channel 201. For example,gigabit-per-second transfer of data can be accomplished using fourtransmission channels, each with one twisted pair cable. Further,transmission channels such as transmission channel 201 can bebi-directional, i.e., transmit data in both directions. For example,receiver 206 may be associated with a transmitter that transmits symbolstreams to other receivers coupled to the same cable as is included intransmission channel 201. Any number of transmitters and receivers maybe coupled to the cable associated with transmission channel 201. Eachcoupling may affect the response of transmission channel 201.

The transmitted symbols in the sequence {a_(k)} are members of thesymbol alphabet {A}. In the exemplary case of PAM-5 signaling, thesymbol alphabet {A} is given by {−2, −1, 0, +1, +2}. The index k againrepresents the time index for each transmitted symbol, i.e., at sampletime k, the symbol being transmitted to transmission channel 201 isgiven by a_(k).

The real-time output of transmitter 221 can be represented as A_(s)(ω),where A_(s)(ω) is the Fourier transform of the analog signal a_(s)(t)that represents the symbol stream {a_(k)}. Therefore,

$\begin{matrix}{{A_{s}(\omega)} = {\int_{- \infty}^{\infty}{{a_{s}(t)}{\mathbb{e}}^{{- {\mathbb{i}\omega}}\; t}\ {{\mathbb{d}t}.}}}} & (2)\end{matrix}$Signal a_(s)(t) also represents the effects of any pre-shaping that maybe performed by transmitter 221.

The output signal y_(k) or Y_(s)(ω) from transmission channel 201, nowtreated as an analog signal, suffers from channel distortion, theaddition of random noise, and a flat signal loss. Referring to FIG. 2A,channel output signal Y_(s)(ω) is the Fourier transform of the analogsignal that represents the output signal stream {Y_(k)} fromtransmission channel 201. Channel output signal y_(k) or Y_(s)(ω) isinput to receiver 206.

As shown in FIG. 2B, transmission channel 201 can be modeled as having alinear, time invariant portion 202 with transfer function H_(s)(ω) and anoise portion represented as noise adder 203. The transfer functionH_(s)(ω) includes the effects of transmit and receive transformers andthe transmission medium (e.g., cable) on the transmitted signal. Theinput signal A_(s)(ω) to transmission channel 201 and thus to portion202 is related to the output signal X_(s)(ω) of portion 202 by therelationshipX _(s)(ω)=H _(s)(ω)A _(s)(ω).  (3)The total output signal Y_(s)(ω) from transmission channel 201 then is,Y _(s)(ω)=H _(s)(ω)A _(s)(ω)+n _(s)(ω),  (4)where n_(s)(ω) is a random noise component. Equations 3 and 4 assume alinear, time invariant transmission system.

For long cable lengths, the intersymbol interference contained in signalY_(s)(ω) can be severe, including significant portions of previouslytransmitted symbols in Y_(s)(ω). For example, at a cable length of aboveabout 100 meters, the contribution of the last sent symbol to thecurrently received signal may be as high as 95%.

Receiver system 206 contains an analog amplifier 222, a pre-filter 207,and a receiver 208 constituted as a digital filter. Amplifier 222amplifies signal y_(k) or Y_(s)(ω) from transmission channel 201.Pre-filter 207 is described in the immediately following paragraphs.Digital filter 208 contains an anti-aliasing filter 209, ananalog-to-digital converter 210, a digital amplifier 211, a digitalequalizer 212, a slicer 213, a coefficient update 214, a digitalautomatic gain control 215, a clock recovery 216, a phase detector 217,and an analog automatic gain control 220. Similar to how components 217and 220 are depicted in FIG. 6B below, components 217 and 220 could bedescribed as outside digital filter 208 since their output signals go tocomponents that precede digital filter 208.

Pre-filter 207 receives the amplified signal from amplifier 222 andpre-shapes that signal for input to digital filter (receiver) 208. Thepre-shaping performed by pre-filter 207 can include partial removal ofintersymbol interference so that less intersymbol 20 interferenceremains to be removed by digital equalizer 212.

Pre-filter 207 can be designed based on frequency-sampling methods inwhich a desired frequency response is uniformly sampled and filtercoefficients are then determined from these samples using an inversediscrete Fourier transform. For example, one embodiment of pre-filter207 includes a one-zero two-pole filter having a frequency response ofapproximately the inverse of, for example, the transfer functionH_(s)(ω) associated with a 50 meter cable (CAT-5) in combination withany pre-shaping that may have been performed by transmitter 221.Pre-filter 207, therefore, can be fixed to remove the influence ofintersymbol interference from a given cable configuration, e.g., atwisted-copper pair having a particular length. Variations in theintersymbol interference inherent in variations of the cable or itslength from that expected can be accommodated by adaptive functions indigital equalizer 212.

Although pre-filter 207 can be any number of filters coupled in series,pre-filter 207 can be represented with a transfer function H_(PF)(ω)that represents the effects on an input signal of all of the filters inpre-filter 207. Therefore, assuming that pre-filter 207 is linear andtime-invariant, the Fourier transform output signal Z_(s)(ω) frompre-filter 207 is given byZ _(s)(ω)=GH _(PF)(ω)Y _(s)(ω),  (5)where G is the analog gain of analog amplifier 222. The transferfunction H_(T)(ω) that represents the combination of transmissionchannel 201, amplifier 222, and pre-filter 207 is given byH _(T)(ω)=GH _(s)(ω)H _(PF)(ω).  (6)Ideally, if pre-filter 207 completely compensates for transmissionchannel 201, the total transfer function H_(T)(ω) is unity. In apractical transmission system, the transfer function H_(PF)(ω) ofpre-filter 207 is determined by inverting the predicted or measuredtransfer function H_(s)(ω) of transmission channel 201.

The frequency response H_(c)(f, 1) of the complete channel, i.e.,transmission system 200 including transmission channel 201 and digitalfilter 208, neglecting random noise n_(s)(ω) and not including thefrequency response of pre-filter 207, can be modeled asH _(c)(f,1)=H _(PR)(z)H _(s)(f,1)H _(EQ)(z)gG H _(co)(f)  (7)where H_(PR) (z) is the partial response shaping accomplished bytransmitter 221 before transmission, z equals e^(jωT), ω equals 2πf, fis the frequency, and T is the symbol (and sampling) interval.H_(s)(f,1) is the frequency response of transmission channel 201, e.g.,of the CAT-5 cable, of length l and the transmit and receivetransformers. In one embodiment, the partial response shaping H_(PR) (z)equals 0.75+0.25z⁻¹ where z⁻¹ represents a one-symbol period delay.H_(EQ)(z) is the transfer function of digital equalizer 212 and isgenerally given by

${\sum\limits_{k = {- N}}^{M}\;{c_{k}Z^{- k}}},$where N and M are positive integers. Inasmuch as z⁻¹ represents aone-symbol period delay, z represents a one-symbol period advance. Inone embodiment, equalizer transfer function H_(EQ) (z) is chosen to bec⁻¹z+c₀+c₁z⁻¹. The parameter g is the output gain of automatic gaincontrol (AGC) 215 in digital filter 208. H_(co)(f) represents thefrequency response of the remaining elements of the complete channel,e.g., analog-to-digital converter 210 (whose pulse can be a rectangularpulse of length T or a trapezoidal pulse with rising and falling edgesof length T/2 and flat portion of length T/2) and other elements oftransmission channel 201.

The frequency response H_(s) (f,1) of transmission channel 201 is afunction of cable length l. Both gain g and digital equalizer transferfunction H_(EQ) (z) will depend on cable length l. The gain g isincreased for increased cable length l due to increased signal loss. Thecoefficient parameters c⁻¹, c₀, and c₁ of equalizer transfer functionH_(EQ) (z) also depend on cable length l. Channel-remainder frequencyresponse H_(co)(f) is not a function of cable length l.

Examples of the frequency response for transmission system 200 are shownin FIGS. 3A through 3D. FIG. 3A shows the transfer function H_(s) (ω) oftransmission channel 201. As shown in FIG. 3A, transmission-channeltransfer function H_(s) (ω) approaches zero asymptotically. FIG. 3Bshows signal e^(−sτ) where s equal jω, and τ is the timing phasedifference discussed below. FIG. 3C represents the transfer functionH_(PF)(s) of pre-filter 207. FIG. 3D represents the total frequencyresponse or the product of the signals represented in FIGS. 3A, 3B and3C.

The frequency response H_(c)(f,1) of the complete channel does notinclude the effects of pre-filter 207. The transfer function H_(PF) (s)of analog pre-filter 207 can be represented by (b₁s+l)/(a₂s²+a₁s+1),where s again equals jω. Pre-filter transfer function H_(PF) (s),therefore, is characterized by the filter parameters b₁, a₁, and a₂.Transfer function H_(PF)(s) can be determined by minimizing a costfunction that is related to the total intersymbol interference found intransmission system 200.

A measure E(1) of the intersymbol interference due to the comparison ofthe folded spectrum with a flat spectrum can be expressed asE(l)=∫_(−1/2T) ^(1/2T) |[H _(c)(f,l)H _(PF)(s)e ^(jωτ)]_(fold)−1|²df  (8)

The parameter τ is the timing phase difference between the transmitterdigital-to-analog converter (not shown) and the receiveranalog-to-digital converter (ADC 210), as calculated by clock recovery216. The integral in Equation (8) represents the inverse discreteFourier transform of all signals received in one period, e.g., −0.5/T to0.5/T. The folded spectrum in the integral can be described by spectrumfolding, which can be defined as

$\begin{matrix}{{\left\lbrack {X(f)} \right\rbrack_{fold} = {\frac{1}{T}{\sum\limits_{k}\;{{X\left( {f - \frac{k}{T}} \right)}.}}}},} & (9)\end{matrix}$where X(f) is any general function of frequency.

In one embodiment, the transfer function H_(PF)(s) of analog pre-filter207 is obtained by minimizing the cost function C given as

$\begin{matrix}{C = {{\sum\limits_{k = 1}^{K}\;{w_{k}{E\left( l_{k} \right)}}} + {w_{K + 1}P}}} & (10)\end{matrix}$with respect to the filter parameters b₁, a₁, and a₂where w_(i) is aweight factor, l_(i) is the ith cable length, K is the number of cablelengths, and P is a high frequency penalty. The first K terms are ameasure of intersymbol interference at cable lengths l₁, l₂, . . .l_(K). In one embodiment, K equals 3. Although any number K of cablelengths can generally be used, minimizing cost function C for K equal to1 results in an implementation of pre-filter 207 optimized for only onecable length. Alternatively, using too many cable lengths complicatesthe optimization. The last term P in Equation (10),P=∫ _(1/2T) ^(∞) |H _(PF)(s)|² df  (11)imposes an additional penalty on the high frequency components ofpre-filter transfer function H_(PF)(s). The high frequency penalty Poperates to attenuate high frequency echoes. Other factors can beincluded in a cost function. For example, a term to reduce quantizationnoise can be added. This quantization term would be proportional to g√c₁²+c₂ ²+ . . . +c_(K) ².

Each term in the cost shown in Equation (10) is weighted by a weightfactor w_(i). These weights specify the importance of each term. Theweights are chosen such that the peak magnitude of pre-filter transferfunction H_(PF)(s) is not too large and so that transfer functionH_(PF)(s) is small at high frequencies. The analog pre-filter 207determined by transfer function H_(PF)(s) found by optimizing costfunction C of Equation 10 minimizes the intersymbol interference forcable lengths l₁ through l_(K) and attenuates high frequency echosignals.

As previously described, transmission-channel transfer functionH_(s)(ω,l), gain g, and equalizer transfer function H_(EQ)(z) all dependon cable length l. Timing phase difference τ from clock recovery 216also depends on cable length l. Therefore, intersymbol interferencemeasures E(l₁) through E(l_(K)) are all different. The parameters G, g,τ, the equalizer parameters in equalizer transfer function H_(EQ)(z)(e.g., c⁻¹, c₀, and c₁), and the measurement parameters in intersymbolinterference measures E(l₁) through E(l_(K)) are those parameters thatthe adaptive loops in analog gain control 220, gain control 215, clockrecovery 216, and coefficient update 214 converge for cable lengthsthrough l_(K), respectively.

Minimizing intersymbol interference measure E(l) with respect toparameters b₁, a₁, and a₂ should enable transfer function H_(PF)(s) forpre-filter 207 to produce a flat folded spectrum if the cable lengthis 1. However, this is based on the assumption that the actual equalizerparameters for equalizer transfer function H_(EQ)(z), analog gain G,digital gain g, and timing phase τ are the same as those used inEquation 8 for measure E(l). If they are different, the results are lessuseful.

The better determination of equalizer parameters for equalizer transferfunction H_(EQ)(z), gain g, and timing phase τ is found by an iterativeprocedure as described below, resulting in determination of pre-filtertransfer function H_(PF)(s). With an initial choice of equalizerparameters for equalizer transfer function H_(EQ)(z), gain g, and timingphase τ, the cost function C is minimized to determine an initialversion of pre-filter transfer function H_(PF)(s). Using this H_(PF)(s)version, the equalizer parameters for equalizer transfer functionH_(EQ)(z), gain g, and timing phase τ are determined for each cablelength l₁ through l_(K). Using these new sets of equalizer parametersfor transfer function H_(EQ)(z), gain g, and timing phase τ (one set ofparameters for each cable length l₁ through l_(K)) in the cost functionC, pre-filter transfer function H_(PF)(s) is recomputed. This process isrepeated until there are no significant changes between successiveiterations. In other words, the above procedure converges to aparticular set of filter parameters for transfer function H_(PF)(s) thatdetermines pre-filter 207.

In one case, transmission-channel transfer function H_(s)(ω) includesthe frequency response of the transmit and receive transformers, each ofwhich is modeled as a first order transfer function with −3 dB cutoff at100 MHz. Additionally, transmission channel 201 is a category-5 twistedcopper pair cable, equalizer transfer function H_(EQ)(z) equalsc⁻¹z+c₀+c₁z⁻¹, partial response shaping H_(EQ)(z) equals 0.75+0.25z⁻¹,and pulse length T equals 8 ns. The optimization of the cost function Cin Equation 10 with K equals to 3 and cable lengths l₁ equals to 0 m,l₂equal to 50 m, l₃ equal to 120 m leads to filter transfer functionH_(PF)(s) for pre-filter 207 described by

$\begin{matrix}{{{H_{PF}(s)} = {\frac{{0.8077\hat{s}} + 1}{{0.1174{\hat{s}}^{2}} + {0.1255\hat{s}} + 1}.}},} & (12)\end{matrix}$where ŝequals sT.

Alternatively, pre-filter 207 can be an adaptive analog filter. Transferfunction H_(PF)(s) for an adaptive analog version of pre-filter 207 canbe of the formH _(PF)(s)=(1−V _(c))+V _(c)PF(s)  (13)and is controlled by the single parameter V_(c)where PF(s) is an analogfilter function. The parameter V_(c) is varied in the range 0<V_(c)<1 toachieve partial equalization for various cable lengths. If V_(c) equals0, pre-filter transfer function H_(PF)(s) is 1 (unity), i.e., noequalization is performed by pre-filter 207. If V_(c) equals 1, transferfunction H_(PF)(s) is analog filter function PF(s), i.e., maximumattainable equalization is achieved attainable by the filter structuredefined by analog function PF(s) for pre-filter 207. As V_(c) is variedlinearly from 0 to 1, pre-filter transfer function H_(PF)(s) varies fromunity to analog function PF(s).

Analog filter function PF(s) can represent a band-pass or high-passfilter. Therefore, the peak magnitude of the frequency response ofpre-filter transfer function H_(PF)(s) increases with increasing V_(c).If analog function PF(s) performs suitable equalization for a particularcable length l_(o), pre-filter 207 with V_(c)<1 performs suitably forcable length l<l_(o). Hence V_(c) is monotonic with cable length l.

For example, analog filter function PF(s) can have one zero and twopoles (complex-conjugate pair) in the form

$\begin{matrix}{{{{PF}(s)} = {\frac{\omega_{n}^{2}}{\omega_{z}} \cdot \frac{s + \omega_{z}}{s^{2} + {2{\delta\omega}_{n}s} + \omega_{n}^{2}}}},} & (14)\end{matrix}$where ω_(z) is the zero frequency, ω_(n) is the pole frequency, and δ isa damping factor.

At low frequency, the filter described by Equation 14 starts from unityand rolls off as 1/s at high frequencies. Hence the filter passes lessnoise and high frequency echo. Moreover, a small order PF(s) requiresfewer resistors, capacitors, and operational amplifiers to realize thecircuit, which implies less sources of circuit noise and also easier andcheaper implementation for pre-filter 207. In another embodiment, analogfilter function PF(s) is the optimized analog filter function thatoptimizes the cost C described in Equation 10 for one cable length wherethat length is the maximum targeted cable length. Parameter V_(c) can beadapted, then, to shorter cable lengths.

To minimize the peak magnitude of the filter structure H_(PF)(s), twostages of filter structures, namely pre-filter transfer functionH_(PF)(s) equals H_(PF)(s)=H₁(s)H₂(s) where H₁(s) and H₂(s) are therespective transfer functions for a pair of cascaded analog filters, canbe utilized. In this case,H ₁(s)=(1−V _(c1))+V _(c1)PF(s),  (15)andH ₂(s)=(1−V _(c2))+V _(c2)PF(s).  (16)For example, analog filter function PF(s) could be a one-zero two-polesfilter with the zero at 30 MHz and complex-conjugate pair poles at 70MHz with a damping factor of 0.4. That is, zero frquency ω_(z) equals60π×10⁶ radians/sec., pole frequency ω_(n) equals 140π×10⁶ radians/sec.,and damping factor δ equals 0.4 in Equation 14 above. A cascade offilter transfer functions H₁(s) and H₂(s) each with the above PF(s)analog function can provide good partial equalization for a wide rangeof cable lengths.

In one embodiment, the digital equalizer transfer function H_(EQ)(z)executed by equalizer 212 can be expressed in the formH _(EQ)(z)=c ⁻¹ z+c ₀ +c ₁ z ⁻¹ +c ₂ z ⁻² + . . . +c _(K) z ^(−K)  (17)The first two coefficients c⁻¹ and c₀ can be fixed (i.e., coefficientupdate 214 does not alter coefficient c⁻¹ or c₀). For example, the firsttwo equalizer coefficients can be set at c⁻¹ equal to −1/8 and c₀ equalto 1. The remaining equalizer coefficients c₁ through c_(K) areadaptively chosen by coefficient update 214. The parameter K can be anypositive integer. For a fixed (non-adaptive) analog filter, equalizercoefficient c₁ decreases monotonically with cable length. Therefore,equalizer coefficient c₁ is a good indicator of cable length.Additionally, AGC gain g is also a good indicator of cable length.Equalizer coefficient c₁ or gain g can be compared to a thresholdTh_(AEQ) and the result of that comparison used to adapt analogpre-filter 207.

In FIG. 2A, phase detector 217 executes an updating algorithm withequalizer coefficient c₁ in order to choose adaptive parameters foranalog pre-filter 207. In phase detector 217, a phase detectionparameter PD_(AEQ) can be calculated byPD_(AEQ)=−(c _(1 −Th) _(AEQ)).The amount of threshold TH_(AEQ) determines how much equalization isperformed in analog pre-filter 207 and how much is performed in digitalequalizer 212. In one example, coefficient c₁ varies between about −0.35to about −1.0 and threshold TH_(AEQ) is chosen to be about −0.4.

Phase detector 217 operates to control pre-filter parameter V_(c). In acascading prefilter, phase detector 217 controls any number of adaptiveanalog filter parameters V_(c1) through V_(cN) where N is the totalnumber of cascaded analog prefilters included in analog pre-filter 207.One method of adaptively choosing a value for parameter V_(c) (or eachof parameters V_(c1) through V_(cN)) is to increment or decrement thevalue of V_(c) based on whether the calculated phase detection parameterPD_(AEQ) is positive or negative. Alternatively, phase detector 217 mayinclude an accumulator that inputs the calculated parameter PD_(AEQ) andoutputs a signal that controls parameter V_(c).

Additionally, in receiver (digital filter) 208 of FIG. 2A, analog AGC220 and analog amplifier 222 scale the input signal to analog pre-filter207, and thus input signal Z_(s)(ω) to digital filter 208, so that theentire dynamic range of ADC 210 is utilized while keeping theprobability of saturation very low. Analog AGC 220 inputs the signaloutput of ADC 210.

Analog AGC 220 outputs a signal to amplifier 222 which adjusts theoutput level of pre-filter 207 to optimize the functionality of ADC 210.In one embodiment, AGC 220 calculates a phase detector parameterPD_(AGC) for the loop, accumulates the results of the phase detectparameter calculation, and converts the accumulated phase detectorparameter to an analog signal which is input by pre-filter 207. Phasedetector parameter PD_(AGC) for this loop can be defined asPD_(AGC) [k]=α _(k,1)+α_(k,2),  (19)where

$\begin{matrix}{\alpha_{k,1} = \left\{ {\begin{matrix}{{- 1}\mspace{14mu}} & {{{if}\mspace{14mu}{\alpha_{k}}} > {Th}_{AGC}} \\0 & {otherwise}\end{matrix},} \right.} & (20) \\{\alpha_{k,2} = \left\{ {\begin{matrix}{1\mspace{14mu}} & {{{if}\mspace{14mu} k\;{{mod}N}} = 0} \\0 & {otherwise}\end{matrix},} \right.} & (21)\end{matrix}$Variable α_(k) is the output signal from ADC 210 during time period k,i.e., at time instant t equal to kT, and modulus number N is chosen tomake use of the range of ADC 210.

At the convergence of the phase loop in AGC 220, i.e., the steady-statecondition, the expected value of PD_(AGC) is 0. This ensures that theprobability of |α_(k)| being greater than Th_(AGC) is 1/N for any timeperiod k. The threshold value Th_(AGC) and modulus number N are suitablychosen to make good use of the A/D range. For the application of GigabitEthernet, Th_(AGC) and N are chosen such that the probability ofsaturation of ADC 210 is less than about 10⁻⁶. In one example, Th_(AGC)is about 0.8 of the range of ADC 210, for example, 50 in a 7 bit ADC,and N is about 1024.

In general, pre-filter 207 can be arranged to reduce or eliminate theintersymbol interference inherent in any length cable. Once a transferfunction, such as that given in Equation 12 or 14, is determined for aparticular configuration of transmission channel 201, one skilled in theart of filter design can construct the appropriate filter. Therefore, atransfer function such as that shown in Equation 12 or 14 completelydescribes an analog filter which can be utilized for equalization inpre-filter 207.

As shown in FIG. 2A, the analog signal output Z_(s)(ω) , which is theinput symbol sequence {a_(k)} distorted by the transmission channel andfiltered by pre-filter 207 in the above described fashion, is input todigital receiver 208. Anti-aliasing filter 209 receives analog signalZ_(s)(ω) from analog pre-filter 207. In most embodiments, anti-aliasingfilter 209 is an analog low pass filter.

Analog-to-digital converter 210 is coupled to receive an output signalfrom anti-aliasing filter 209. ADC 210 can have any accuracy, but inmost embodiments a six to eight bit converter is utilized. Due topre-filter 207, the linearity, i.e., number of bits, requirement of ADC210 is reduced. For example, by using a 50-meter cable (CAT-5) plustransmit shaping, as described above, the ADC requirements can besignificantly reduced if receiver 206 includes a pre-filter implementingthe transfer function described by Equation 8. The requirements of ADC210 may be reduced from an 8-bit ADC to a 6-bit ADC at 125 megasamplesper second, for example.

By reducing the linearity of the ADC requirements, a linear equalizer isused in one embodiment rather than a decision feedback equalizer or amore complicated trellis decoder. In addition, by using pre-filter 207,critical timing loops normally associated with Gigabit receiver designsare eliminated. Experiment has shown that the time complexity of thecritical path required to implement a 4D, 8-state trellis decoder in aGigabit receiver is reduced. The reduction in complexity inherent inreducing the distortion in the signal input to digital receiver 208 canresult in receivers having fewer components and simpler implementations.

A discrete-time model of the response of transmission channel 201 incombination with pre-filter 207 is shown in FIG. 4 and includes achannel response 204, represented by the channel function f(z), and anoise adder 205. Noise adder 205 represents addition of a random noisefactor n_(k) to the transmitted signal. The discrete-time model isparticularly applicable for digital receiver 208. In that case, transferfunction f(z) is a folded spectrum of the combined frequency responseH_(PR)(z)H_(s)(ω)H_(PF)(ω)H_(co)(ω).

It is assumed that the channel model includes the effect of transmit andreceive filtering. In addition, the transmission channel 201 is assumedto be linear in that two overlapping signals simply add as a linearsuperposition. Therefore, the channel function polynomial f(z) ofchannel response 204 can be defined asf(z)=f ₀ +f ₁ z ⁻¹ +f ₂ z ⁻² + . . . +f _(N) z ^(−N),  (22)where f₀, . . . , f_(j), . . . , and f_(N) are the polynomialcoefficients representing the dispersed component of the (k−j)th symbolpresent in the a_(k)th symbol a_(k), z⁻¹ represents a one-symbol perioddelay, and N is a cut-off integer such that f_(j) for j>N is negligible.The polynomial f(z) represents the frequency response of transmissionchannel 201 in combination with pre-filter 207. See A. V. OPPENHEIM & R.W. SCHAFER, DISCRETE-TIME SIGNAL PROCESSING 1989.

The noiseless output signal x_(x) of transmission channel 201 at sampletime k, i.e., the output signal from channel response 204, is then givenbyx _(k) =f ₀ *a _(k) +f ₁ *a _(k−1) + . . . f _(N) *a _(k−N)  (23)Thus, the channel output signal at time k depends, not only ontransmitted data at time k, but also on past values of the transmitteddata, i.e., there remains some intersymbol interference.

The noise element of the input signal, represented by noise adder 205,is represented by the sequence {n_(k)}. Therefore, the noisy output ofthe channel, i.e., the output signal from ADC 210, is given byα_(k) =x _(k) +n _(k),  (24)where the noise samples (n_(k)) are assumed to be independent andidentically distributed Gaussian random variables (See E. A. LEE AND D.G. MESSERCHMITT, DIGITAL COMMUNICATIONS (1988)) with variance equal toσ².

Digital amplifier 211 amplifies the output signal a_(k) fromanalog-to-digital converter 210 to adjust for the loss of signalresulting from the transmission through transmission channel 201 andpre-filter 207. Equalizer 212 equalizes the amplified version of signalα_(k) to produce equalized signal a_(k)′ as indicated in FIG. 2A.

Equalizer 212 can be any type of equalizer including a linear equalizer,a decision feedback equalizer, or a sequence detector, alone or incombination. Examples of equalizers applicable to 100 or 1000 BASE-TEthernet over category-5 wiring, 24 gauge twisted copper pair, arediscussed in U.S. patent application Ser. No. 08/974,450, filed Nov. 20,1997, Raghavan, assigned to the same assignee as the presentapplication, now U.S. Pat. No. 6,083,269, herein incorporated byreference in its entirety; and U.S. patent application Ser. No.09/020,628, filed Feb. 9, 1998, Raghavan, assigned to the same assigneeas the present application, now U.S. Pat. No. 6,115,418, hereinincorporated by reference in its entirety.

Further examples of equalization systems are described in U.S. patentapplication Ser. No. 09/296,086, filed Apr. 21, 1999, Raghavan et al.,assigned to the same assignee as the present application, now U.S. Pat.No. 6,418,172 B1, herein incorporated by reference in its entirety; U.S.patent application Ser. No. 09/151,525, filed Sep. 11, 1998, Raghavan,assigned to the same assignee as the present application, now U.S. Pat.No. 6,415,003 B1, herein incorporated by reference in its entirety; U.S.patent application Ser. No. 09/161,346, filed Sep. 25, 1998, Raghavan etal., assigned to the same assignee as the present application, now U.S.Pat. No. 6,438,163 B1, herein incorporated by reference in its entirety;and U.S. patent application Ser. No. 09/560,109, filed Apr. 28, 2000,Sallaway et al., assigned to the same assignee as the presentapplication, now U.S. Pat. No. 7,050,517 B1, herein incorporated byreference in its entirety.

Slicer 213 receives signal stream {a_(k)′} from equalizer 212 and, basedon that stream {a_(k)′}, decides on an output symbol stream {â_(k)}. Theoutput symbol stream {â_(k)} represents the best estimate of receiver208 of the symbol stream {a_(k)} that was originally transmitted bytransmitter 221.

Receiver 208 may be an adaptive receiver, further including coefficientupdate 214 that adjusts the coefficient parameters of equalizer 212 inorder to optimize the performance of receiver 208. Receiver 208 may alsoinclude automatic gain control (AGC) 215 that dynamically adjusts thegain of amplifier 211 in order to maximize the efficiency of receiver208. Furthermore, clock recovery 216 can provide timing and framing foranalog-to-digital converter 210, representing an element of aphase-locked loop.

FIG. 5A shows an embodiment of another receiver 506 according to thepresent invention. Receiver 506 includes pre-filter 207, anti-aliasingfilter 209, analog-to-digital converter 210, amplifier 211, and digitalequalizer 212. Although digital equalizer 212 can be any equalizersystem, as has been previously described, digital equalizer 212 in FIG.5A is shown as having an equalizer 511 coupled in series with a trellisdecoder 512. Equalizer systems are described in U.S. patent applicationSer. Nos. 08/974,450, 09/020,628, 09/161,346, 09/296,086, 09/151,525,and 09/560,109, all cited above, will not be further discussed here.

Receiver 506 also includes adaptive coefficient update 214 whichadaptively chooses the operating parameters of equalizer 511, gaincontrol 215 which adaptively chooses the gain setting of amplifier 211,and clock recovery 216 which forms the phase-locked-loop required toframe the data acquisition by analog-to-digital converter 210.

Receiver 506 can further include a baseline wander correction circuit510 that, when combined with adder 515, corrects the output signal α_(k)of analog-to-digital converter 210 for signal wander. Baseline wandercorrection is further described in U.S. patent application Ser. No.09/151,525, cited above. Receiver 506 can also include an A/D referenceadjuster 517, which adjusts the reference voltage of analog-to-digitalconverter 210 according to the measured apparent length of the cableassociated with transmission channel 201.

Receiver 506 can include a cable quality and length calculator 518. Asdescribed in U.S. patent application Ser. No. 09/161,346, cited above,cable quality and length calculator 518 calculates the length of cablein transmission channel 201 and the quality of transmission channel 201based on the gain calculation of gain control 215 or the equalizercoefficients of equalizer 511. Both A/D reference adjuster 517 and cablequality and length calculator 518 are affected by pre-filter 207, whichhas the effect of simultaneously making transmission channel 201 appearto be of very high quality and to make the cable length of transmissionchannel 201 appear longer. The apparent quality increases becausepre-filter 207 removes some of the interference caused by transmissionchannel 201. The cable appears longer if there is any loss of signalstrength in pre-filter 207. Cable quality and length calculator 518 can,however, adjust for the presence of pre-filter 207 in order to haveaccurate calculations of cable length and quality.

Receiver 506 can also include an echo canceller 513 and a NEXT canceller514. NEXT canceller 514 cancels interference on one transmission linebased on the transmission of symbols over neighboring lines. Echocanceller 513 cancels interference from symbols transmitted by atransmitter (not shown) associated with receiver 506.

In some transmission systems, signals are transmitted over a cablehaving multiple wires. Transmission channel 201 and receiver 506represent detection of the transmitted signal over one of the multiplewires. In that case, signals on neighboring wires affect the transmittedsignal on transmission channel 201. NEXT canceller 514 computes theinfluence of transmitted signal from other pairs of wires at the inputof adder 519. The projected influence from symbols transmitted onneighboring lines is subtracted from the digitized symbol by adder 519.

Echo canceller 513 subtracts the influence of symbols that are reflectedback into receiver 506 by transmission along a cable associated withtransmission channel 201. In most transceiver systems, receiver 506 anda transmitter (not shown) are coupled to a common host. The transmittertransmits signals through transmission channel 201 to a receivercounterpart (not shown) of transmitter 221. Some of that transmittedsignal may be reflected back into receiver 506. Echo canceller 513projects the reflected signal based on the transmitted signals andsubtracts the influence of that signal at adder 516 and adder 519.

FIG. 5B shows an embodiment of pre-filter 207 that is sensitive to thecable length of transmission channel 201. Pre-filter 207 as shown inFIG. 5B includes pre-filters 520-1 through 520-N. Pre-filters 520-1through 520-N execute transfer functions H_(PF) ¹(ω) through H_(PF)^(N)(ω), respectively. Each of pre-filters 520-1 through 520-N isoptimized to counter the interference from a transmission channel havinga particular cable length. Each pre-filter 520-i can be designed byminimizing a cost function such as that shown in Equation 10. A selector521, in response to the cable length L calculated by cable quality andlength calculator 518 (FIG. 5A), selects one of pre-filters 520-1through 520-N. Selector 521 controls a switch 522 which supplies inputsignal Y_(s)(ω) to the selected one of pre-filters 520-1 through 520-N.Therefore, pre-filter 207 can be selected in order to optimize theperformance of receiver 506.

FIG. 5C shows another embodiment of pre-filter 207 according to thepresent invention. Pre-filter 207 of FIG. 5C executes the adaptivelycontrolled transfer function of Equation 13. Input signal Y_(s)(ω) isinput to block 525, which executes the transfer function PF(s). Transferfunction PF(s) can, for example, be the transfer function of Equation14. The input signal Y_(s)(ω) is also input to block 526 which executesthe transfer function one. The output signal from block 525 ismultiplied by the adaptively chosen parameter V_(c) in multiplier 527and input to adder 529. The output signal from block 526 is multipliedby 1−V_(c) in multiplier 218 and added to the output signal frommultiplier 527 by adder 529. The output signal from adder 529 is theoutput signal Z_(s)(ω) from pre-filter 207.

FIG. 6A shows a multi-wire receiver 600 according to the presentinvention. Transmission receiver 600 receives input analog signalstreams {y_(k) ⁽¹⁾} through {y_(k) ^((M))} from M wires 603-1 through603-M, respectively. Signal streams {Y_(k) ⁽¹⁾} through {Y_(k) ^((M))}are also indicated as input signals Y_(s) ⁽¹⁾(ω) through Y_(s)^((M))(ω), respectively, in FIG. 6A. Each receiver input signal Y_(s)^((i))(ω) is the Fourier transform of receiver input analog signal y_(k)^((i)) for integer i running from 1 to M. Each of input signals Y_(s)⁽¹⁾(ω) through Y_(s) ^((M))(ω) includes the effects of a transmissionchannel 601, as described above for transmission channel 201.Additionally, each of signals Y_(s) ⁽¹⁾(ω) through Y_(s) ^((M))(ω)includes effects of cross talk between wires so that, for example,signal Y_(s) ^((i))(ω), where the ith wire 603-i is an arbitrary one ofwires 603-1 through 603-M, includes a contribution from signals on allof the other wires, i.e., wires 603-1 through 603-(i−1) and wires603-(i+1) through 603-M.

Individual receivers 602-1 through 602-M receive input signals Y_(s)⁽¹⁾(ω) through Y_(s) ^((M))(ω), respectively, i.e., input analog signalstreams {y_(k) ⁽¹⁾} through {y_(k) ^((M))}, respectively, and generateoutput signal streams {a'_(k) ⁽¹⁾} through {a'_(k) ^((M))},respectively. In some embodiments, signal streams {a′_(k) ⁽¹⁾} through{a′_(k) ^((M))}are input to slicers (not shown in FIG. 6A) withinreceivers 602-1 through 602-M, respectively. The slicers in receivers602-1 through 602-M, determine symbol streams {â′_(k) ⁽¹⁾} through{â′_(k) ^((M))}(also not shown in FIG. 6A), respectively, as discussedbelow in connection with FIG. 6B for an embodiment of one of receivers602-1 through 602-M. Symbol streams {á_(k) ⁽¹⁾} through {á_(k) ^((M))}here are temporary decisions made in order to control the adaptation ofparameters within receivers 602-1 through 602-M, respectively.

An arbitrary receiver 602-i, which is one of receivers 602-1 through602-M, also inputs the output symbol streams, {Tx_(k) ⁽¹⁾} through{Tx_(k) ^((M))}, from a transmitter 606 associated with receiver 600.Each of receivers 602-1 through 602-M can then include echo cancellationand near end crosstalk (NEXT) cancellation due to the transmittedsymbols of transmitter 606. As indicated in FIG. 6A, each output symbolstream {Tx_(k) ^((i))} is also supplied on corresponding wire 603-i totransmission channel 601.

In some embodiments, receiver output signal streams {a′_(k) ⁽¹⁾} through{a′_(k) ^((M))} are input to a delay skew compensator 604. FIG. 6Adepicts such an embodiment. Delay skew compensator 604 provides outputsignal streams, also denoted as {a′_(k) ⁽¹⁾} through {a′_(k) ^((M))}here, that are input to a multi-dimensional (M-D) decoder 605 for finaldecision on the received symbols.

Delay skew compensator 604 aligns the M signal streams {a′_(k) ⁽¹⁾}through {a′_(k) ^((M))} , i.e., compensator 604 aligns signals a′_(k)⁽¹⁾ through a′_(k) ^((M)) at each time period (or clock cycle) k, sothat any delays between signal streams {a′_(k) ⁽¹⁾} through {a′_(k)^((M))} received from receivers 602-1 through 602-M, respectively, areremoved. Relative delays between signal streams {a′_(k) ⁽¹⁾} through{a′_(k) ^((M))} may be introduced in transmission channel 601 or byreceivers 602-1 through 602-M. The aligned signals a′_(k) ⁽¹⁾ througha′_(k) ^((M)) from delay skew compensator 604 for particular clock cyclek arrive at M-D decoder 605 simultaneously.

Decoder 605, which may be a Viterbi decoder, uses aligned signal streamsa′_(k) ⁽¹⁾ through a′_(k) ^((M)) to make a final decision on theincoming data. The final decision of decoder 605 is indicated in FIG. 6Aas M output symbol streams â′_(k) ⁽¹⁾ through â′_(k) ^((M)) .

Additionally, decoder 605 may utilize an error detecting code such asthat defined in the IEEE standard for Gigabit Ethernet. See, e.g., IEEE802.3ab, “Gigabit Long Haul Copper Physical Layer Standards Committee”,1997 Standard. In one embodiment, M-D decoder 605 is a Viterbi decoderwhich makes a final decision on data which has been encoded by an8-state Ungerboeck code, as described in the IEEE Gigabit Spec. TheViterbi decoder in this embodiment is a maximum likelihood sequenceestimator, as described in Viterbi, A.J., “Error Bounds forConvolutional Codes and an Asymptotically Optimum Decoding Algorithm,”IEEE Trans. Inf Theory, IT-13, pages 260-269, April 1967, hereinincorporated by reference in its entirety. M-D decoder 605, therefore,maximizes the probability of correctly estimating the entire sequence ofsymbols.

FIG. 6B shows an embodiment of a receiver 602-i that includes an analogprefilter 619 and a digital filter 620. Analog prefilter 619 includes aDC offset adder 610 coupled to a DC offset correction circuit 628, anecho canceller adder 611 coupled to an analog echo canceller circuit627, an analog multiplier 612 coupled to analog automatic gain controlcircuit 220, and analog equalizers 613 and 614 coupled to analogequalizer adaptor circuit (phase detector) 217. Digital filter 620includes digital equalizer 212, a digital echo/NEXT canceller adder 615coupled to a digital echo canceller 621 and to NEXT cancellers 618-1through 618-M without a canceller 618-i, AGC booster (digital amplifier)211 coupled to digital automatic gain control circuit 215, a baselinewander substracter (or adder) 616 coupled to a baseline wandercorrection circuit 617, and slicer 213. Analog portion 619 is coupled todigital portion 620 through analog-to-digital converter 210. Forexemplary purposes, slicer 213 is shown as a PAM-5 decoder. Timingrecovery loop (clock recovery) 216 controls a clock used in both theanalog and digital portions of receiver 602-i and calculates the timingphase parameter τ_(k) ^((i)).

Slicer 213 provides a temporary decision â_(k) ^((i)) on the kth symbola_(k) ^((i)) intended to be transmitted in signal stream {y_(k) ^((i))}on wire 603-i and (b) an error e_(k) ^((i)) based on input signal a'_(k)^((i)), where error e_(k) ^((i)) is defined ase _(k) ^((i)) =a' _(k) ^((i)) −â _(k) ^((i)).  (25)The temporary decision â_(k) ^((i)) and error e_(k) ^((i)) are utilizedin various circuit loops in receiver 602-i in order to adapt parametersin receiver 602-i. As discussed below and indicated in FIG. 6B,temporary decision â_(k) ^((i)) and error e_(k) ^((i)) are also utilizedto adapt parameters in analog prefilter 619.

DC offset correction circuit 628 includes an ADCO control 633 coupled toa digital-to-analog converter (DAC) 634. DAC 634 provides a signal whichis negatively added to the received signal Y_(s) ^((i))(ω), in DC offsetadder 610. ADCO control 633 inputs the output signal α_(k) ^((i)) fromADC 210 and estimates the DC offset that occurs in analog prefilter 619.This calculated DC offset, upon being converted from digital to analogfrom by DAC 634, is then subtracted from the input signal Y_(s)^((i))(ω) in adder 610.

Analog echo canceller circuit 627 includes an AEC control 629, DACs 630and 631, and an RC circuit 632. AEC control 629 inputs the error signale_(k) ^((i)) as well as the transmitted symbol stream {Tx_(k) ^((i))} onwire 603-i (FIG. 6A), and adapts the resistance R_(k) ^((i)) andcapacitance C_(k) ^((i)) in RC circuit 632. Transmit signal Tx_(k)^((i)) is filtered in RC circuit 632. Echo adder 611 substracts theresultant filtered signal from input signal Y_(s) ^((i))(ω) minus the DCoffset determined by DC offset correction circuit 628. The parametersR_(k) ^((i)) and C_(k) ^((i)) are adapted to approximately duplicate theeffects of the transmit signal Tx_(k) ^((i)) on the signal input toadder 611. Appropriate values for R_(k) ^((i)) and C_(k) ^((i)) minimizethe residual echo from the transmit signal Tx_(k) ^((i)), which resultsin minimizing the requirements of digital echo canceller circuit 621.Furthermore, by minimizing the residual echo, analog AGC 220 can providefor maximum boost to input signal Y_(s) ^((i))(ω)through multiplier 612without overloading ADC 210, which results in clipping. The additionalboost at multiplier 612 results in a lessened need for amplification atdigital AGC booster 211, thereby minimizing quantization noise.

Analogous to what occurs in receiver 206 of FIG. 2A, analog gain controlcircuit 220 outputs a gain signal to multiplier 612 that adjusts theoutput levels of prefilter 619 to optimize the functionality of ADC 210.Analog gain control circuit 220 contains AGC control circuit 625 and DAC626. FIGS. 6C and 6D show embodiments of analog gain control circuit220, specifically embodiments of AGC control circuit 625.

One embodiment of an AGC control circuit 625 is shown in FIG. 6C. Testblock 630 compares signal α_(k) ^((i)) with threshold TH_(AGC) andcalculates, for each receiver 602-i, value α_(k,1) according to Equation20. The value α_(k,2) is calculated according to Equation 21. The valueof PD_(AGC) is calculated according to Equation 19 in adder 631. Thevalue of PD_(AGC) is input to an adder 632. The output signal of adder632 goes to saturation block 633. Saturation block 633 saturates at, forexample, 13 bits. The output from saturation block 633 is delayed oneclock cycle and added to PD_(AGC) at adder 632. The combination of adder632, saturation block 633, and delay 635 forms an accumulator.

The output signal of saturation block 633 is right shifted by aparticular number of bits, for example 7 bits, in shifter 634 to give anoutput signal of a particular number of bits, for example, 6 bits. Theoutput signal from shifter 634 provides an input signal to DAC 626.Multiplier 612 multiplies the analog output signal from DAC 626, whichis the output signal from analog AGC control circuit 220 (FIG. 6B), ismultiplied by the input signal Y_(s) ^((i))(ω) as modified by thesubtractions at adders 610 and 611.

Because of the low frequency nature, the input signal to DAC 626 of AGC220 has very small variations from sample to sample. In most cases, thevariation is at most one count. FIG. 6D shows an embodiment of analogAGC control circuit 220 (AGC control 625 and DAC 626) that takesadvantage of this feature. Instead of a “general purpose” 6 bit D/A, aless expensive Sigma-Delta D/A is used for DAC 626 in the embodiment ofFIG. 6D. In that case, saturation block 633 is replaced with a smallerblock 636 of size, for example, 7 bits. The accumulated value, i.e., theoutput signal from the accumulator formed by adder 632, block 636 anddelay 635, is wrapped around (modulo) to a particular number of bits,for example 7 bits.

The output of block 636 is a three-level signal representing overflow,no change, or underflow of the accumulation value. The three-levelsignal is the output signal received by DAC 626 implemented as aSigma-Delta DAC. DAC 626 then outputs an analog value which multiplier612 multiples by the input signal Y_(s) ^((i))(ω) again as modified bythe subtractions at adders 610 and 611.

In the embodiment shown in FIG. 6B, the analog equalization isaccomplished by analog equalizer 613 cascaded with analog equalizer 614.Each of analog equalizer 613 and analog equalizer 614 is controlled byanalog equalizer control circuit (phase detector) 217. Analog equalizercontrol circuit 217 includes AEQ control 622 coupled to DAC 623, whichis coupled to control analog equalizer 613, and coupled to DAC 624,which is coupled to control analog equalizer 614. Analog equalizers 613and 614 accomplish partial equalization of the input signal Y_(s)^((i))(ω), resulting in a lessened requirement for digital equalization.Analog equalizers 613 and 614 and analog equalization control circuit217 operate as is described above in connection with Equations 14through 18.

Analog-to-digital converter 210 receives the output signal Z_(s)^((i))(ω) from analog prefilter 619 and digitizes the signal. The outputfrom ADC 210 is signal α_(k) ^((i)).

ADC 210 samples input signal Z_(s) ^((i))(ω) based upon the clock outputfrom timing recovery loop (clock recovery) 216 and phase τ_(k) ^((i)).Clock recovery 216 recovers the frequency of the received signal (i.e.,the frequency of transmitter 221 (FIG. 2)) and finds the optimal value τof the timing phase τ_(k) ^((i)) of the incoming signal. For a constantclock frequency offset between the remote transmitter'sdigital-to-analog converter and ADC 210, the optimal timing phase τvaries linearly with time. The rate of change of phase τ_(k) ^((i)) isproportional to the clock frequency offset.

Clock recovery 216 can be a second order loop. One embodiment of clockrecovery 216 is shown in FIG. 6E. A phase detector 650 estimates thedifference between the optimal phase τand the current value of τ_(k)^((i)) based on the output symbol â_(k) ^((i)) and the error calculatione_(k) ^((i)). The output signal PD_(CR) from phase detector 650 forreceiver 602-i can be determined in several manners, including a slopemethod and a Mueller & Muller (M&M) method. In the M&M method, theoutput signal PD_(CR) from phase detector 650 isPD_(CR) =e _(k−1) ^((i)) â _(k) ^((i)) −e _(k) ^((i)) â _(k−1) ^((i)).  (26)

In the slope method,PD_(CR) =e _(k) ^((i))slope(k),  (27)where

$\begin{matrix}{{{slope}(k)} = \left( {\begin{matrix}{1,} & {{{if}\mspace{14mu}{\hat{a}}_{k - 1}^{(i)}} < {\hat{a}}_{k}^{(i)} < {\hat{a}}_{k + 1}^{(i)}} \\{{- 1},} & {{{if}\mspace{14mu}{\hat{a}}_{k - 1}^{(i)}} > {\hat{a}}_{k}^{(i)} > {\hat{a}}_{k + 1}^{(i)}} \\{0,} & {otherwise}\end{matrix}.} \right.} & (28)\end{matrix}$

The output signal PD_(CR) from the phase detector 650 is input to a loopfilter 651 that has a proportional part and an integral part. The outputsignal from loop filter 651, indicating the correction on the clockfrequency, is input to a frequency controlled oscillator 652 whichcauses ADC 210 to sample at an optimal phase by controlling the samplingfrequency of ADC 210. Frequency controlled oscillator 652, in otherwords, outputs a clock signal whose zero-crossings are given by NT+τ_(k)^((i)).

If the coefficient c⁻¹ of digital equalizer 212 is adapted, theadaptation algorithms between coefficient update 214 and clock recovery216 will interact adversely, often causing failure of receiver 600. Toprevent this interaction, coefficient c⁻¹ is fixed, for example, at −⅛,in order that the timing loop can converge to an optimum phase.

Since part of the equalization is accomplished in analog equalizers 613and 614, digital equalizer 212 can be simplified. For example, digitalequalizer 212 can be a linear equalizer without causing large amounts ofnoise enhancement. Of course, as has been previously discussed, otherembodiments of digital equalizer 212 can use any equalization scheme.

High frequency signals are attenuated more by transmission channel 601than are low frequency signals. The equalization, between analogequalizers 613 and 614 and digital equalizer 212, then should equalizethe attenuation difference across the frequency band.

In one embodiment, digital equalizer 212 in each receiver 602-i 212 is alinear equalizer executing the transfer function H^((i)) _(EQ)(z) givenas

$\begin{matrix}{H_{EQ}^{(i)} = {{c_{k,{- 1}}^{(i)}z} + c_{k,0}^{(i)} + {c_{k,1}^{(i)}z^{- 1}} + \ldots + {c_{k,K}^{(i)}{z^{- K}.}}}} & (29)\end{matrix}$The parameter K can be any positive integer, for example, in someembodiments. The coefficient c^((i)) _(k, −1) can be fixed, for example,at −⅛, to avoid interaction with the adaptation performed by timingrecovery loop 216. Further, the coefficient c^((i)) _(k,0) can be fixed,for example, at 1, to avoid interaction with digital AGC 215. Theremaining equalizer coefficient c^((i)) _(k,1) through c^((i)) _(k,K)are adaptively chosen by coefficient update 214. Equalizer transferfunction H^((i)) _(EQ)(z) of Equation 29 corresponds to equalizertransfer function H_(EQ)(z) of Equation 17 with each coefficient c^((i))_(k,j) of Equation 29 replacing corresponding coefficient c_(j) ofEquation 17.

Coefficient update 214 can use a least mean squares (LMS) technique tocontinuously adjust the equalizer coefficients c^((i)) _(k,j) such thatC ^((i)) _(k+1,j=c) ^((i)) _(k,j)−μ^((i)) _(EQ, j)sign(α^((i)) _(k−j)e)^((i)) _(k).   (30)

The LMS technique minimizes the mean squared error, which is a functionof intersymbol interference and random noise, of the input signal atslicer 213. The parameter μ_(EQ,j) ^((i)) controls the rate at which thecoefficient c_(k,j) ^((i)) changes. In some embodiments, the parameterμ_(EQ,j) ^((i)) is set to about 10⁻³ on chip powerup and reduced toabout 10⁻⁵ for continuous operation.

After equalization with digital equalizer 212, digital echo canceller621 removes the residual echo due to transmitter 606 transmitting onwire 603-i which is left by analog echo canceller circuit 627. The M-1NEXT cancellers 618-1 through 618-M remove the near end crosstalk (NEXT)from transmitter 606 on wires 603-1 through 603-M, respectively, otherthan wire 603-i. In a four-wire system (M=4), there are three NEXTcancellers 618-1 through 618-M except for 618-i and one echo canceller621 for signals transmitted on wire 603-i.

Digital echo canceller 621 cancels the residual echo not cancelled byanalog echo canceller circuit 627. The bulk of the echo cancellation isaccomplished by analog echo canceller circuit 627. Removing the residualecho by digital echo canceller 621 is necessary to achieve the bit-errorrate (BER) performance of receiver 602-i.

In one embodiment, echo canceller 621 uses a finite-impulse response(FIR) filter to estimate the residual echo on the channel. FIR echocanceller 621 executes a transfer function EC^((i)) _(k) given by

$\begin{matrix}{{{EC}_{k}^{(i)} = {\sum\limits_{p = 0}^{L}\;{ϛ_{k,p}^{(i)}z^{- p}}}},} & (31)\end{matrix}$where L is an integer, for example, 64 and 56. Echo canceller 621 inputsthe transmitted symbol stream {Tx_(k) ^((i))} and estimates the residualecho at that point in the data path, including the impulse response ofthe residual echo channel after analog echo canceller 627, analog AGC625, analog equalizers 613 and 614, and digital equalizer 212.

Each of the coefficients ζ^((i)) _(k,j) in Equation 31 is chosen by anadaptation loop using a least mean squares technique such thatζ^((i)) _(k+1,j=ζ) ^((i)) _(k,j−μ) ^((i)) _(EC,jsign() Tx ^((i))_(k−j)e) ^((i)) _(k).   (32)The coefficients ζ_(k+1,j) ^((i)) are continuously adjusted to maintainthe minimum mean squared error at slicer 213. The parameter μ_(EC,j)^((i)) may initially be set high (e.g., 10⁻³) and then lowered (e.g.,10⁻⁵) for continuous operation.

As mentioned above, the M-1 NEXT cancellers 618-1 through 618-M inreceiver 602-i cancel the near end crosstalk which is a result oftransmitter 606 transmitting on wires 603-1 through 603-M other thanwire 603-1. Note that there is no NEXT canceller (618-i ) for receiver602-i because the effects of transmitting symbols on wire 603-i arecancelled by analog echo canceller 627 and digital echo canceller 621.Each of the M-1 NEXT cancellers 618-1 through 618-M estimates theimpulse response from the NEXT in an FIR block. The impulse responsethat is used to estimate the NEXT at this point in the data path is theimpulse response of the NEXT contribution in transmission channel 601that has been added to the receive signal filtered by analog prefilter619 and digital equalizer 212. Each of the M-1 NEXT cancellers 618-1through 618-M executes a transfer function NE^((i)) _(p,k) given by

$\begin{matrix}{{{NE}_{p,k}^{(i)} = {\sum\limits_{l = 0}^{L}\;{\xi_{p,k,l}^{(i)}z^{- l}}}},} & (33)\end{matrix}$where p denotes a channel that is not channel i and L can be anypositive integer, for example, 44 or 16.

Each of the coefficients ξ^((i)) _(p,k,l) is adaptively chosen accordingto a least mean squares technique such thatξ^((i)) _(p,k+1,j=ξ) ^((i)) _(p,k,j+μ) ^((i)) _(NE,p,j)sign(Tx ^((p))_(k−j))e ^((i)) _(k).   (34).

The coefficients ξ^((i)) _(p,k,j) are continuously updated to maintainthe minimum mean squared error at slicer 213. The parameter μ^((i))_(NE,p,j)may initially be set high (e.g., ˜10⁻³) and then lowered (e.g.,˜10⁻⁵) for steady state operation.

The echo and NEXT estimations performed by echo canceller 621 and theM-1 NEXT cancellers 618-1 through 618-M are subtracted from the outputsignal of equalizer 212 by adder 615.

Digital AGC 215 inputs a gain signal g_(k) ^((i)) to AGC booster 211which digitally amplifies the output signal from adder 615. The signalis boosted by AGC booster 211 to levels determined by slicer 213. Thegain g_(k) ^((i)) is set to counter the losses resulting fromtransmission channel 601 and not recovered in analog prefilter 619.During acquisition, the gain g_(k) ^((i)) can be updated by the equationg ^((i)) _(k+1=g) ^((i)) _(k−μ) ^((i)) _(AGC)(e ^((i)) _(non,k)).   (35)with error e^((i)) _(non,k) determined frome ^((i)) _(non,k=|a') ^((i)) _(k)(i)|−TH^((i)) _(AGC).   (36)

where TH^((i)) _(AGC) is the average absolute value of a'^((i)) _(k).The parameter μ^((i)) _(AGC) can initially be set high and then loweredduring steady state operation. During steady state operation, a leastmean squares approach can be taken, in which caseg ^((i)) _(k+1=g) ^((i)) _(k−μ) ^((i)) _(AGC)sign(â ^((i)) _(k))e ^((i))_(k).   (37)

Finally, baseline wander correction circuit 617, in combination withbaseline wander subtracter 616, corrects for baseline wander. Adiscussion of baseline wander can be found in U.S. patent applicationSer. No. 09/151,525, cited above.

One skilled in the art will recognize that the components of receiver506 may be arranged differently. For example, in FIG. 6B amplifier 211follows equalizer 212 while in FIG. 2A equalizer 212 follows amplifier211. One skilled in the art will also recognize that receivers accordingto the present invention may not have some of the features shown inFIGS. 2A, 5A, and 6B or, alternatively, may have other features notshown in FIGS. 2A, 5A, and 6B. FIGS. 2A, 5A, and 6B, therefore, are notexhaustive of all configurations of receivers that are nonethelesswithin the scope of this disclosure.

The above examples, therefore, are demonstrative only. One skilled inthe art can recognize variations which fall within the scope of thisinvention. As such, the invention is limited only by the followingclaims.

1. A receiver system comprising: a pre-filtering analog equalizer foradaptively performing analog equalization on asymbol-information-carrying input analog signal affected withintersymbol interference, or on a first intermediate analog signalgenerated from the input analog signal, to produce a filtered partiallyequalized analog signal with reduced intersymbol interference; ananalog-to-digital converter for converting the filtered analog signal,or a second intermediate analog signal generated from the filteredanalog signal, into an initial digital signal; a digital equalizer foradaptively performing digital equalization on the initial digitalsignal, or on an intermediate digital signal generated from the initialdigital signal, to produce an equalized digital signal as a stream ofequalized digital values with further reduced intersymbol interference;an output decoder for decoding the equalized digital values, orintermediate digital values generated from the equalized digital values,into a stream of symbols; and equalization control circuitry foradjusting equalization filter characteristics of the analog and digitalequalizers such that adjustments of the filter characteristics of aspecified one of the equalizers depend adaptively on adaptiveadjustments of the filter characteristics of the other of theequalizers.
 2. A receiver system as in claim 1 wherein the controlcircuitry adaptively adjusts the filter characteristics of the specifiedequalizer in response to information provided by operating on theequalized digital values or on further digital values generated from theequalized digital values.
 3. A receiver system as in claim 1 wherein thecontrol circuitry adaptively adjusts (a) a specified parameter of thefilter characteristics of the specified equalizer in response to anerror signal generated by decoding the equalized digital values orfurther digital values generated from the equalized digital values and(b) a specified parameter of the other of the equalizers in response tothe adaptively adjusted parameter of the filter characteristics of thespecified equalizer.
 4. A receiver system as in claim 3 wherein theerror signal at any time during operation of the receiver system variesaccording to the difference between (i) the equalized digital value atthat time or a further digital value generated from that equalizeddigital value and (ii) the value of a corresponding one of an alphabetof predefined symbols from which the stream of symbols is substantiallyformed, the corresponding predefined symbol being produced by decodingthat equalized digital value or that further digital value.
 5. Areceiver system as in claim 4 wherein the predefined symbols used ingenerating the error signal are generated along a different signalprocessing path than the stream of symbols.
 6. A receiver system as inclaim 3 further including an additional decoder for generating the errorsignal.
 7. A receiver system as in claim 6 wherein the additionaldecoder comprises a slicer.
 8. A receiver system as in claim 1 furtherincluding an echo canceller for causing the equalized digital values orthe intermediate digital values generated from the equalized digitalvalues to be produced with reduced effects of echo in the analog inputsignal.
 9. A receiver system as in claim 8 wherein the echo cancellerperforms echo reduction on at least one of (a1) the input analog signal,(a2) a third intermediate analog signal generated from the input analogsignal, (a3) the filtered analog signal, and (a4) a fourth intermediateanalog signal generated from the filtered analog signal or/and on atleast one of (b1) the initial digital signal, (b2) a furtherintermediate digital signal generated from the initial digital signal,(b3) the equalized digital values, and (b4) further digital valuesgenerated from the equalized digital values.
 10. A receiver system as inclaim 8 wherein the echo canceller performs echo reduction on at leastone of (a1) the input analog signal, (a2) a third intermediate analogsignal generated from the input analog signal, (a3) the filtered analogsignal, and (a4) a fourth intermediate analog signal generated from thefiltered analog signal and on at least one of (b1) the initial digitalsignal, (b2) a further intermediate digital signal generated from theinitial digital signal, (b3) the equalized digital values, and (b4)further digital values generated from the equalized digital values. 11.A receiver system as in claim 10 wherein the echo canceller adaptivelyperforms echo reduction.
 12. A receiver system as in claim 8 wherein theecho canceller operates in response to error information indicative ofhow much the equalized digital values or further digital valuesgenerated from the equalized digital values differ from decodedinformation generated by decoding the equalized or further digitalvalues.
 13. A receiver system as in claim 12 wherein the decodedinformation utilized in generating the error information is generatedalong a different signal processing path than the stream of symbols. 14.A receiver system as in claim 1 further including a crosstalk cancellerfor causing the equalized or/and intermediate digital values to beproduced with reduced effects of crosstalk in the input analog signal.15. A receiver system as in claim 14 wherein the crosstalk cancellerperforms the crosstalk reduction at least partially adaptively.
 16. Amethod comprising: adaptively performing analog equalization on asymbol-information-carrying input analog signal affected withintersymbol interference, or on a first intermediate analog signalgenerated from the input analog signal, to produce a filtered partiallyequalized analog signal with reduced intersymbol interference;converting the filtered analog signal, or a second intermediate analogsignal generated from the filtered analog signal, into an initialdigital signal; adaptively performing digital equalization on theinitial digital signal, or on an intermediate signal generated from theinitial digital signal, to produce an equalized digital signal as astream of equalized digital values with further reduced intersymbolinterference; decoding the equalized digital values, or intermediatedigital values generated from the equalized digital values, into astream of symbols; and adjusting equalization filter characteristicsduring the equalization-performing acts such that adjustments of thefilter characteristics during a specified one of theequalization-performing acts depend adaptively on adaptive adjustmentsof the filter characteristics during the other of theequalization-performing acts.
 17. A method as in claim 16 wherein thecharacteristics-adjusting act comprises adaptively adjusting the filtercharacteristics during the specified equalization-performing act inresponse to information provided by operating on the equalized digitalvalues or on further digital values generated from the equalized digitalvalues.
 18. A method as in claim 16 wherein thecharacteristics-adjusting act comprises adaptively adjusting (a) aspecified parameter of the filter characteristics during the specifiedequalization-performing act in response to an error signal and (b) aspecified parameter of the filter characteristics during the other ofthe equalization-performing acts in response to the adaptively adjustedparameter of the filter characteristics during the specifiedequalization-performing act.
 19. A method as in claim 18 wherein theadaptively adjusting act includes: decoding one of the equalized digitalvalues, or a further digital value generated from that digital value, toproduce a corresponding one of an alphabet of predefined symbols fromwhich the stream of symbols is substantially formed; and generating theerror signal to vary according to the difference between (i) thatequalized digital value or that further digital value and (ii) the valueof the corresponding predefined symbol.
 20. A method as in claim 19wherein the predefined symbols used in generating the error signal aregenerated along a different signal processing path than the stream ofsymbols.